1. Technical Field
The present invention relates generally to electronic systems that employ a phase control technique to control the amount of power delivered from an AC line source to a load. The present invention more specifically relates to using a modified Costas loop that incorporates rectangular to polar conversion in its feedback leg to recover the fundamental frequency phase information of an AC power line without detecting a zero crossing.
2. Background Art
Lighting control systems typically perform a dimming function by altering the conduction angle of a switchable conductive device, such as a metal oxide field effect transistor (MOSFET), by delivering a signal to a gate of the MOSFET such that the timing of the signal varies with the selected dimming level.
FIG. 1 shows an alternating current (AC) fundamental 11 sinusoidal source voltage such as 60 Hz, 120 VAC as is commonly used in the United States. Those skilled in the art will recognize that such a sinusoidal source voltage includes positive peaks 113, negative peaks 114 and both positive trending zero crossings 111 and negative trending zero crossings 112. These zero crossings are typically used as timing references for controlling the conduction angle used by typical light dimming equipment.
In a typical forward phase control system, generation of a gate control signal is synchronized with the AC fundamental 11 such that, sometime after a zero crossing of the AC line voltage is detected, the gate control signal is generated, the gate of the switchable conductive device is energized, and the device conducts for the remainder of the AC half cycle. During the time interval between the detection of the zero crossing and the generation of the gate control signal, no power is delivered from the AC source to the load (i.e. the device is non-conducting). The conducting and non-conducting time intervals are altered in response to a dimming command. Ference (U.S. Pat. No. 5,430,356) provides a description of such programmable lighting control that includes normalized dimming for different light sources.
Also shown in FIG. 1 is a noise 12 signal represented by an out of phase sawtooth wave at a third harmonic (e.g. 180 Hz) of AC fundamental 11. When the noise 12 is overlaid on the AC fundamental 11, a representative noisy AC line 13 signal is formed. The noise 12 shown may be representative of certain interfering signals encountered in ‘real-world’ lighting applications.
Prior art has detected noisy AC line zero crossings 131 using a phase lock loop (PLL) to reduce the jitter caused by interfering signals, such as noise 12. Newman (U.S. Pat. No. 6,091,205) teaches the use of a Bessel low-pass filter before a zero crossing detector, but disadvantageously relies on either factory calibration or analysis of a lightly filtered version of the AC line to compensate for the phase delay introduced by its Bessel low-pass filter. Those skilled in the art will recognize that relying on factory calibration will render the equipment susceptible to errors due to component drift over time, errors due to operational temperature variations, and increased production cost during manufacture. Also since the Newman method relies on monitoring a ‘lightly filtered’ zero cross, such factory calibration may prove to be of little value if the AC line was noisy at the time when that calibration was performed.
Newman also discloses several other filters that may be used to filter the incoming signal, including a Butterworth filter, and a Chebyshev filter. According to Newman, after the incoming signal is filtered, it is then passed to a zero-crossing detector to start the timing sequence typically used in a phase control dimmer.
In the unrelated field of telecommunications, a Costas loop is a phase-locked loop used for carrier phase recovery from suppressed-carrier modulation signals, such as from double-sideband suppressed carrier signals. The Costas loop includes a ‘quadrature demodulator’, a type of circuit which forms the front end of almost all modern narrow-band radio frequency (RF) receivers from cell-phones to digital cable boxes.
FIG. 2 shows a typical prior art implementation of a Costas loop 20 that involves two parallel tracking loops operating simultaneously from the same reference oscillator 243. The first tracking loop, consists of an in-phase leg (I-leg) and a common feedback leg and operates as phase locked loop (PLL) based on an in-phase (e.g. phase shift=0°) modulation signal from the reference oscillator 243. The second tracking loop, consists of a quadrature leg (Q-leg) and the common feedback leg and operates as phase locked loop (PLL) based on a quadrature (e.g. phase shift=90°) modulation signal from the reference oscillator 243. The quadrature modulation signal is typically produced by passing the reference oscillator output through a 90° phase shift device 244
An I-leg product detector 221 accepts an input signal, such as an RF carrier 21, and the in-phase modulation signal and produces a multiplicative I-leg product signal. The I-leg product signal is passed through an I-leg low pass filter 222 and the resultant filtered product is the I-leg output I(t) signal 225 that is applied to a feedback leg phase detector 241.
A Q-leg product detector 231 accepts the RF carrier 21 and the quadrature modulation and produces a multiplicative Q-leg product signal. The ( ) leg product signal is passed through a Q-leg low pass filter 232 and the resultant filtered product is the Q-leg output Q(t) signal 235 that is applied to the feedback leg phase detector 241.
The feedback leg phase detector 241 multiplies the I-leg output I(t) signal 225 with Q-leg output Q(t) signal 235 and the resultant loop error product signal is passed through a loop filter 242 and then used to modify the output frequency of the reference oscillator 243. The loop error signal should settle to a fixed value when the loop is locked. A negative loop error typically results in a lower reference oscillator 243 frequency, and similarly, a positive loop error typically results in a higher reference oscillator 243 frequency. The low pass filters in each leg must be wide enough to pass the desired modulation, such as data modulation, without distortion.
Some key aspects of a quadrature demodulator, such as a Costas loop 20, are that bandwidth is twice the corner frequency of the low-pass filters 222 and 232, the center frequency of the filter is determined by the reference oscillator 243 frequency, and all spectral information in the band passed by the Costas loop 20 (e.g. filter) is contained in the I(t) and Q(t) outputs 225 and 235.
In the field of lighting control, the prior art has attempted to detect zero crossings by either operating upon a separately generated signal that is intended to replicate both the phase and frequency of the fundamental of the AC line, or by operating upon the AC line itself. Unfortunately, merely quantifying an AC line with a zero crossing comparator throws away useful signal information.
For light dimming applications, the frequency of the AC line is not known exactly, but can be assumed to be within a range of a few Hz from the applicable national standard (i.e. 50 or 60 Hz). In practice, most national power grids are extremely accurate although backup power generators are not. The Costas loop, such as that shown in FIG. 2 is not suitable for recovering phase information from a noisy AC line because the output amplitude of the multiplier feeding the loop filter is highly dependent on the amplitude of the incoming AC signal. If a modified Costas loop is to be adapted for light dimming applications, then some means is needed to adjust the frequency of the reference oscillator to make it track the AC power line.
To solve the aforementioned problems, the present invention is a unique apparatus that synthesizes square waves that are substantially in phase and in quadrature with an AC source where the parameters used to synthesize the square waves are also used to control the conduction angle of the load.